A commonly used principle for transmitting data over a radio channel and for overcoming the signal rate limitation of binary sequence signalling is to make use of four or more unique symbols. Thereby, the bit rate can exceed the maximum signal rate (in bits/s) corresponding to double the pass-band (in Hz) as given by the Nyquist theorem.
Quadrature phase shift keying (QPSK) also denoted 4-state quadrature amplitude modulation (4-QAM) involves that two-bit words are coded into four discrete symbols. These symbols can be represented as signal vectors in the complex plane having constant amplitude but four distinct phase values in relation to a reference signal. Detection is carried out by establishing to which quadrant in the complex plane the received signal can be referred.
If a higher modulation order is used, the bit rate can be increased further. However, higher requirements are inflicted on the detection stage since it becomes more difficult to distinguish the individual symbols from one another, as they appear closer in the complex plane. The deterioration of the signal as transmitted over a given media also constitutes a limitation to the possible number of symbols being used.
Higher order keying is commonly referred to as M'ary QAM, where M=2N refers to the number of discrete symbols being available, whereby N bits can be transmitted per symbol. M'ary QAM is also referred to as M'ary APK (amplitude phase shift keying), as both the amplitude and phase may vary for individual symbols.
FIG. 1 shows a conventional transmitter and FIG. 2 shows a conventional receiver.
The transmitter unit comprises a data buffer 1, a mapper 2, baseband filtering unit 3, intermediate frequency (IF) oscillator 6, phase divider 5, adders 7, and summer 4 from which a radio frequency (RF) signal is transmitted.
Data stored temporarily in buffer 1 is conveyed to the mapper 2 in accordance with the rate data can be transmitted over the radio interface. The data, which can be seen as a binary bit serial string, is partitioned into symbols by the mapper 2 having an I component and Q component in the complex plane as explained above.
The receiver, on the other hand, decodes I and Q components multiplying the incoming signal (RF) with 90 degree phase skewed signals provided by signal oscillator IF12 from divider D11. The signal of IF 12 is typically rendered coherent by means of a carrier recovery PLL (phase locked loop) with the carrier signal from IF 6, such that the RF signal, after being filtered in respective filters 9 and 10, can be decoded back into the complex plane. An error signal 16 corresponding to the deviation of the detected symbol value from an expected symbol value is fed into PLL loop back filter 13 adjusting IF generator IF 12.
FIGS. 3 and 4 show a conventional scheme for transmitting data. A frame alignment word F1 consisting of a predetermined sequence of symbols functions as a reference for subsequent frames of traffic data B1, B2 . . . BN−1. For example, the frame-word may have a length of 8 bits. After transmission of a fixed period of frames, the frame alignment word is repeated. Via a frame-aligner 15, in which the predetermined sequence is recovered, the demodulator, can identify the individual frame position for each frame.
As is shown in FIG. 4′, the frame alignment word may comprise a single pilot signal P, which is discernible from the remaining traffic carrying symbols T.
An error signal vector E corresponding to the deviation of the detected symbol value D from an expected reference symbol value R is detected in de-mapper 14 and is fed into PLL loop back filter 13, which deviates a control value E′, also denoted deviated error signal. For instance the angle φ between vectors for points D and R, can be calculated and used as error control signal E′.
The latter signal is used to adjust IF generator IF 12, so that the phase of signal from IF12 is rendered coherent with the signal of IF6.
Additive noise, which consists mainly of thermal noise in the receiving signal, will typically be transferred to the phase detector output. The noise part of the received signal constitutes a constant area around the transmitted constellation symbol, as the noise part is independent of the given symbol.
All QAM schemes larger than 4 have constellations for which the envelope varies for the individual symbols. Hence, if the error signal E is used directly and unprocessed for QAM schemes larger than 4, the noise transfer from symbols with a small envelope, G, will be much larger than symbols with a large envelope, H. This relation has been indicated in FIG. 3, which discloses detected symbols for a 16QAM constellation under the influence of thermal noise.
Therefore, a need has arisen as to compensate for noise contributions.
The optimum with respect to noise transfer would be to “equalise” the phase detector with regard to the envelope, hence to multiply the detected phase error with the envelope of the signal, as expressed below:E′=φ·|D|  Iwhere D is the detected signal and φ is the angle between the detected signal and the decided symbol reference R (square centre). Please confer FIG. 4.
However, the above calculation requires many programming instructions and is therefore not suitable for some applications.
A deviated signal, which is more easily calculated, is given by the expression:E′=D—Q·R—I−D—I·R—Q  IIwhere D is the detected signal and R is the decided symbol.
In the latter case, the noise will get a “square” dependency related to the envelope of the signal. In FIG. 5 various lines have been shown for given values of control signals E′ for relation II above.
In order to equalise the detected error with regard to noise, the error should be divided by the actual envelope. Hence, the following expression may be used:
                    III        ⁢                  :                                              E          ′                =                                            D_Q              ·              R_I                        -                          D_I              ·              R_Q                                                                                            (                  D_Q                  )                                2                            +                                                (                  D_I                  )                                2                                                        
However, also the two above methods of noise balancing require relatively complex algorithms, which then again require extensive processing power in the receiver stage.
Prior art document U.S. Pat. No. 5,796,786 shows a phase error detection method in which a phase error value for the received data is obtained by subtracting the decided I-channel data and multiplying the sign of the difference by the difference itself and applying a weighting function to the phase error value. This signal is used for phase correction of received data. The weighting function is applied to reduce the wrong detection of a phase error caused by a decision error possibly generated in an adjacent error between symbols.